Damping or resonant peaks in an electric motor which is operated using a converter with an intermediate voltage circuit, by means of a transformer-coupled damping resistance, and a corresponding electric motor

ABSTRACT

In a converter system having an intermediate voltage circuit which operates with a supply network-side input inductor in the step-converter mode or has other input-side inductances, there is a risk of natural system oscillations being formed via discharge capacitances in conjunction with motors. If the motor now has an amplitude/frequency response with a pronounced resonant frequency in the region of such natural system oscillations, then there is a risk of higher voltages occurring at the motor star point (S) than in the motor phases (I, V, W). This is prevented by the present invention by introducing an impedance (Z), in particular at the input to the motor, in order to damp capacitive discharge currents to ground potential, which are caused by system oscillations (f sys ) (excited asymmetrically with respect to ground in the motor phases (U, V, W)) of the converter system (L K , UR, LT, M) in the winding sections.

FIELD OF THE INVENTION

The invention relates to a method for damping resonant peaks at a motorstar point in an electric motor which is operated using an intermediatevoltage circuit converter with an input-side inductance, in particular amains system input indicator (supply network-side input inductor,) andwhich, owing to characteristics of its winding sections, has a frequencyresponse with at least one resonant frequency with respect to groundpotential, and to a corresponding electric motor in which resonant peaksare damped in such a manner.

BACKGROUND OF THE INVENTION

In present-day converter systems having an intermediate voltage circuit,in particular in multi-shaft converter systems of this type, systemoscillations can be formed which are virtually undamped. This isespecially true in converters having an intermediate voltage circuit anda regulated supply in the form of a regulated supply network-sideconverter, which is also referred to as an input converter.

Converters of this type are used for operating electrical machines witha variable supply frequency. Such an intermediate circuit frequencyconverter allows an electric motor, for example a three-phase machinesuch as a synchronous machine, to be operated not only in such a mannerthat it is linked directly to the supply network and hence has a fixedrotation speed, but also such that the fixed supply network can bereplaced by an electronically produced, variable-frequency andvariable-amplitude supply for powering the electrical machine.

The two supply systems, (i.e. the supply network whose amplitude andfrequency are fixed, and the supply system which supplies the electricalmachine with a variable amplitude and frequency), are coupled via a DCvoltage storage device or a DC current storage device in the form ofwhat is referred to as an intermediate circuit. In this case, suchintermediate circuit converters essentially have three centralassemblies:

a supply network-side input converter which can be designed to beunregulated (for example diode bridges) or to be regulated, in whichcase energy can be fed back into the supply network only by using aregulated input converter;

an energy storage device in the intermediate circuit in the form of acapacitor in the case of an intermediate voltage circuit and an inductorin the case of an intermediate current circuit; and

an output-side converter or inverter for supplying the machine, whichgenerally uses a three-phase bridge circuit having six active currentdevices which can be turned off, for example IGBT transistors, toconvert the DC voltage in an intermediate voltage circuit into athree-phase voltage system.

Such a converter system having an intermediate voltage circuit which,inter alia, owing to its very wide frequency and amplitude controlrange, is preferably used for main drives and servo drives in machinetools, robots and production machines, is shown in the illustration inFIG. 1.

The converter UR is connected to a three-phase supply network N viafilter F and an energy-storage inductor whose inductance is L_(K). Theconverter UR has the described converted E, an intermediate voltagecircuit with an energy-storage capacitance C_(ZK), and an outputinverter W. FIG. 1 shows a regulated converter E which is operated suchthat it is controlled by switching components (for example a three-phasebridge circuit composed of IGBT transistors), as a result of which thearrangement experiences excitation A1. The inverter W is likewisecontrolled via further switching components, for example, by means of athree-phase bridge circuit having six IGBT transistors. As a result ofthose switching operations the inverter W experiences excitation A2 ofthe system. The capacitor C_(ZK) in the intermediate voltage circuit isconnected between the positive intermediate circuit rail P600 and thenegative intermediate circuit rail M600. The inverter is connected onthe output side via a line LT, having a protective-ground conductor PEand a shield SM, to a motor M in the form of a three-phase machine.

The fixed-frequency three-phase supply network N now supplies theintermediate circuit capacitor C_(ZK) via the input converter E and viathe filter F and the energy-storage inductor L_(K) by means of theregulated supply, with the input converter E (for example apulse-controlled converter) operating together with the energy-storageinductor L_(K) as a step-up converter. Once current flows into theenergy-storage inductor L_(K), it is connected to the intermediatecircuit and drives the current into the capacitor C_(ZK). Theintermediate circuit voltage may therefore be greater than the peakvalue of the supply network voltage.

This combination effectively represents a DC voltage source. Theinverter W uses this DC voltage in the described manner to form athree-phase voltage system in which case, in contrast to the sinusoidalvoltage of a three-phase generator, the output voltage does not have theprofile of an ideal sinusoidal oscillation, but also has harmonics inaddition to the fundamental, since it is produced electronically via abridge circuit.

However, in addition to the described elements in such an arrangement,it is also necessary to consider parasitic capacitances which assist theformation of system oscillations in such a converter system. Thus, inaddition to the filter F with the discharge capacitance C_(F), the inputconverter E, the inverter W and the motor M also have dischargecapacitances C_(E), C_(W) and C_(M) to ground. Furthermore, there isalso a capacitance C_(PE) in the line LT to the protective-groundconductor PE, and a capacitance C_(SM) in the line LT to the groundedshield SM.

It has now been found that these system oscillations are excited to aparticularly pronounced extent in the converter E. Depending on thecontrol method chosen for the supply, two or three phases of the supplynetwork N are short-circuited, in order to pass current to theenergy-storage inductor L_(K). If all three phases U, V, W areshort-circuited, then either the positive intermediate circuit rail P600or the negative intermediate circuit rail M600 is hard-connected to thestar point of the supply network (generally close to ground potentialdepending on the zero phase-sequence system component). If two phases ofthe supply network N are short-circuited, then the relevant intermediatecircuit rails P600 and M600 are hard-connected to an inductive voltagedivider between the two supply network phases.

Depending on the situation relating to the supply network voltages, thisvoltage is in the vicinity of ground potential (approximately 50-60 V).Since the intermediate circuit capacitance C_(ZK) is generally large(continuous voltage profile), the other intermediate circuit rail is 600V lower or higher and can thus also break down the remaining phase ofthe supply network. In both cases, the intermediate circuit isparticularly severely deflected from its “natural” balanced steady-stateposition (±300 V with respect to ground), which represents aparticularly severe excitation for system oscillation.

With respect to the production of undesirable system oscillations, thefrequency band below 50 to 100 kHz area, which is relevant for theapplication, allows a resonant frequency to be calculated based onconcentrated elements. In this case, the discharge capacitances C_(F) toground in the filter F are generally so large that they do not have afrequency-governing effect. In this case, it can be assumed thatdominant excitation to oscillations takes place upstream of thedescribed capacitances, and that the filter discharge capacitance C_(F)can be ignored.

The resonant frequency f_(res)(sys) of this system, which is referred toas f_(sys) in the following text, is thus given by:

where $\begin{matrix}{{f_{sys} = \frac{1}{2\pi \sqrt{L_{\sum} \cdot C_{\sum}}}}{where}} & (1) \\{L_{\sum} = {L_{K} + L_{F}}} & (2)\end{matrix}$

where L_(K) represents the dominant component and L_(F) the unbalancedinductive elements acting on the converter side in the filter (forexample current-compensated inductors); and

C _(Σ) =C _(E) +C _(W) +C _(PE) +C _(SM) +C _(M)  (3)

This relationship is shown in FIG. 2. In this case, L_(Σ) and C_(Σ) forma passive circuit, which is excited by excitation A and starts tooscillate at its natural resonant frequency f_(sys).

Accordingly, in addition to the shifts with an amplitude of 600 V, forexample, that occur during operation, an additional, undesirableoscillation with an amplitude of up to several hundred volts is alsomodulated onto the voltages of the intermediate circuit rails P600 andM600.

In electric motors M in general, but in particular if they are designedusing field coil technology (for example torque motors), a frequencyresponse with pronounced resonant peaks with respect to ground potentialcan occur if such motors are excited in the common mode with respect toground at all the motor terminals, for example due to the undesirablesystem oscillations described above.

These resonant points can be explained by an unbalanced equivalentcircuit comprising a lattice network circuit K of parasitic elements(inductances L and discharge capacitances C) in the motor winding, as inFIG. 3 which shows the winding section of one phase U of a three-phasemotor M with the three phases U, V, W whose winding sections areelectrically connected to one another at the motor star point S. Theinput voltages of the three-phase current generated by the inverter Ware applied to the outer terminals, which are opposite the star point S,of the respective winding sections.

This applies in particular to motors using field coil technology, inwhich individual four-pole networks in the lattice network K arepossible by virtue of the construction, and essentially correspond to asingle field coil. In field coil technology, the magnetic cores, whichare composed of magnetic steel laminates, have teeth which act as polecores, which are placed onto the prefabricated coils and are connectedas appropriate. The individual inductances L are, as can be seen in FIG.3, electrically connected in series, with each field coil beingcapacitively coupled to the pole core (magnetic steel laminate) on whichthe coil is fit. These respective capacitances are represented asdischarge capacitances C to ground, which are formed by the magneticcore.

However, the described phenomenon can also be explained in the case ofmotors with a different construction (for example using what is referredto as wild winding) by a model of a lattice network K, by this modelrepresenting an equivalent circuit with identical four-pole networks inthe form of LC tuned circuits, whose elements simulate the frequencyresponse. The peak in this case occurs in the region of the star pointS, which is normally not deliberately subjected to voltage loads. If thesystem oscillation of the converter system occurs in the vicinity of anatural motor frequency, then the insulation system to ground can beoverloaded, in particular at the star point S, leading to prematurefailure of the motor M, since, due to resonance, considerably highervoltages can occur at the motor star point than at the motor terminals.

This is true for all voltage levels (low-voltage, medium-voltage andhigh-voltage systems), but particularly when the step-up converterprinciple is used (with the energy-storage inductor L_(K)) on theconverter side UR and a frequency response with pronounced resonantpeaks with respect to ground potential occurs on the other side in themotor M as is the case in motors with a particularly low natural motorfrequency because the natural damping in the motor resulting from eddycurrent losses and hysteresis losses etc. is particularly low.

Similar problems arise repeatedly in the field of electrical machineswhen transient overvoltages occur. The overvoltages are thus limited inorder to avoid flashovers. For example, according to German patentdocument DE-A-38 26 282, a voltage-dependent metal-oxide resistor isconnected in parallel with a coil in order to limit overvoltages. InGerman patent document DE-B-28 34 378, winding sections areshort-circuited in order to damp quadrature-axis field. In a similarway, according to German patent document DE-A-24 33 618, transientovervoltages in a synchronous machine are damped by means ofquadrature-axis field damper bars.

Furthermore, European patent document EP-A-0 117 764 describes howovervoltages which occur due to resonance phenomena can be suppressed byferroelectric insulators between the coil windings. Finally, Europeanpatent document EP-B-0 681 361 addresses the problem of higher-orderharmonic oscillations, which can occur in converters and rectifiersusing power thyristors. The damper winding is in consequence connectedto capacitors in order to form tuned circuits. The tuned circuits have aresonant frequency which is six times as high as the fundamentalfrequency of the synchronous machine. Higher-order harmonic oscillationson a fundamental can thus be absorbed. Nevertheless, the problem ofpossible resonant peaks at the star point S of a motor M still remains.

SUMMARY OF THE INVENTION

It is an object of the present invention to avoid resonant peaks excitedby such system oscillations in an electric motor operated using such aconverter system. This object is achieved by a method for dampingresonant peaks at a motor star point in an electric motor which isoperated using an intermediate voltage circuit converter with aninput-side inductance, in particular a supply network-side inputinductor, and which, owing to characteristics of its winding sections,has a frequency response with at least one resonant frequency withrespect to ground potential, in that an impedance for damping capacitivedischarge currents to ground potential, which are caused in the windingsections, is introduced into all the motor phases leading to a motorstar point. For this purpose, this impedance is designed for capacitivedischarge currents which are caused by system oscillations (which areexcited asymmetrically with respect to ground in the motor phases) ofthe converter system.

It has also been found to be preferred if all the motor phases leadingto a motor star point are routed through a lossy magnetic coupling core.In this case, the losses required for damping can advantageously beproduced by the characteristics of the magnetic material itself.Alternatively, this can be achieved if the coupling core, for example amagnetic core, has a winding which is short-circuited via an impedance.In this case, it has been found to be advantageous, particularly withregard to possible retrofitting of motors, for all the motor phasesleading to a motor star point to be routed through the lossy magneticcore at the input to the motor. If the impedance is a non-reactiveresistance, then this results in a particularly simple andcost-effective implementation.

On the basis of the knowledge that each winding section of the motorforms an LC lattice network, the resistance of the non-reactive resistoris preferably determined by:$R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}$

where L is the inductance and C the discharge capacitance of one latticenetwork element in the LC lattice network structure.

The total inductance of the coupling circuit formed with the magneticcore is preferably given by:

where,${L_{H1} \geq \frac{R}{2\pi \quad f_{0}}},{where},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

with f_(res) being a pronounced resonant frequency in theamplitude/frequency response of the motor.

In this case, the method according to the present invention can also beused for an electric motor having a number of motor star points, inparticular for a linear motor or a torque motor, by carrying out themethod for each motor star point, and with a suitable impedance in eachcase being coupled into all the motor phases leading to a motor starpoint. In this case as well, the impedance can be transformed in at themotor input, in which case it is then sensible to route all the motorphases to all the motor star points through a single coupling core,which must be dimensioned as appropriate. A torque motor is a machinewhich is designed to produce high torques, generally at low rotationspeeds, for example in the form of a brushless synchronous motor with alarge number of poles and permanent-magnet excitation.

If the converter is operated together with a supply network inputinductor in order to provide a supply based on the step-up converterprinciple, then the invention results in significant advantages withregard to undamped system oscillations.

Furthermore, the object of the present invention is also achieved by anelectric motor using an intermediate voltage circuit converter having aninput-side inductance, in particular a supply network input inductor,having a frequency response, which is governed by characteristics of itswinding sections with at least one resonant frequency with respect toground potential, and in which all the motor phases leading to a motorstar point are routed through a lossy magnetic coupling core. This canbe easily achieved by using the characteristics of the magnetic materialitself to produce the losses required for damping, or by the magneticcore having a winding which is short-circuited via an impedance. Itpreferred for the magnetic core to be arranged at the input to themotor. Since this impedance is designed for damping capacitive dischargecurrents with respect to ground potential which are caused by systemoscillations (excited asymmetrically with respect to ground in the motorphases) of the converter system in the winding sections, systemoscillations of the converter system can be suppressed particularlyeffectively, together with resonant peaks produced by them in the motor,on the basis of a passive circuit, particularly in conjunction with asupply network-side input inductor. It is particularly preferred for theimpedance to be a non-reactive resistance. The value of thisnon-reactive resistance and of the total inductance is preferablygoverned by the same rules as for the method according to the invention,where$R_{a} \geq {{\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}\quad {and}\quad L_{H1}} \geq {\frac{R}{2\pi \quad f_{0}}.}$

The success of the invention can in this case be improved further by thecoupling core being constructed such that it does not enter saturationat any operating point of the motor. The solution proposed by thepresent invention has been found to be particularly advantageous formotors with winding sections using field core technology, which eachform a lattice network structure composed of inductances L and dischargecapacitances C, with the impedance being used for transformer damping ofthese lattice network structures. This is achieved particularly well ifthe impedance is designed such that it damps common-mode currents, whichare excited asymmetrically with respect to ground in the motor phases,of the converter system in the lattice network structure.

However, the principle of the invention can also be applied to any otherdesired forms of electric motors, particularly also those using what isreferred to as wild winding technology, in particular low-voltagemotors. This has been found to be particularly advantageous for suchdrives whose geometric dimensions are large and in which large slotareas result in large discharge capacitances, which lead to particularlylow resonant frequencies f_(res). This is because the risk of resonantpeaks at the motor star point is low provided such pronounced resonantpoints of the motor are well above any possible converter systemoscillations. However, the situation changes, the closer such resonantfrequencies in the frequency response of a motor with respect to groundpotential are in the region of such converter system oscillations. Thisrelates primarily to the physical size of the motor. The size of a motorgoverns the slot area which itself affects the capacitance C_(M) of themotor with respect to ground potential, since this discharge capacitanceincreases with the size of the slot area. As the discharge capacitanceC_(M) of the motor increases, the pronounced resonant frequency f_(res)in the amplitude/frequency response of the motor with respect to groundpotential in turn falls and thus comes closer to the region ofundesirable natural system frequencies f_(sys) of the converter system.This means that, as the geometric dimensions of the motor, for examplethe physical length or the diameter, increase, pronounced resonantfrequencies come closer to this critical region, and the problem ofresonant peaks becomes more severe.

The present invention effectively counteracts this by means of themeasures described above by providing a means for changing the frequencyresponse of the motor with respect to ground potential such that thereare now virtually no pronounced resonant peaks f_(res) in the vicinityof the natural system frequencies f_(sys) of the converter system shownin FIG. 1.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details and advantages of the invention will be apparent fromthe following description of an exemplary embodiment and in conjunctionwith the drawings, wherein elements having the same functionality aredenoted by the same reference symbols. In the figures:

FIG. 1 shows a block diagram of a converter system having a three-phasemotor using a converter with an intermediate voltage circuit and acontrolled input converter, and a supply network input inductor in thestep-up converter mode;

FIG. 2 shows an equivalent circuit of the passive circuit, formed by thearrangement of a converter system shown in FIG. 1, with regard to systemoscillations;

FIG. 3 shows an outline sketch of a lattice network structure formed ina motor;

FIG. 4 shows a block diagram system model of the effective path of thevoltages with respect to ground potential from the supply network to themotor star point;

FIG. 5 shows a schematic block diagram of a topology in a convertersystem;

FIG. 6 shows an outline sketch of a balanced drive for the motorcomprising the intermediate voltage circuit on the basis of two phasesL1 and L2;

FIG. 7 shows a timing diagram of the voltage profile between these twophases L1 and L2, compared to the switching states of the inverter W;

FIG. 8 shows a timing diagram of the voltage profile of the phase L1with respect to ground;

FIG. 9 shows a corresponding timing diagram of the voltage profile ofthe phase L2 with respect to ground;

FIG. 10 shows an outline sketch of an unbalanced drive for the motorcomprising the intermediate voltage circuit as a common-mode system foranalysis of phase to ground;

FIG. 11 shows a timing diagram of the unbalanced voltage profile of thephases L1 and L2 with respect to ground;

FIG. 12 shows a corresponding timing diagram of the DC component of theunbalanced voltage profile of the phases L1 and L2 with respect toground;

FIG. 13 shows a corresponding timing diagram of the AC component of theunbalanced voltage profile of the phases L1 and L2 with respect toground;

FIG. 14 shows an amplitude/frequency response of a motor with respect toground in order to illustrate the transfer function H₂(s);

FIG. 15 shows an amplitude/frequency response of a motor with respect toground ignoring the natural damping, which increases as the frequencyrises, in order to illustrate the transfer function H₂(s);

FIG. 16 shows an outline sketch of a lattice network structure formedfrom four-pole networks;

FIG. 17 shows an example of a winding layout of a motor winding usingfield coil technology;

FIG. 18 shows a cross-sectional view of the installed position of thesefield coils in the laminated core;

FIG. 19 shows the unbalanced equivalent circuit of such an arrangementas shown in FIG. 17 and FIG. 18;

FIG. 20 shows the same lattice network structure as that in FIG. 3 withthe transformer-coupled damping resistance according to the invention;

FIG. 21 shows an equivalent circuit for the coupling core and theimpedance as shown in FIG. 20; and

FIG. 22 shows a comparison of the amplitude/frequency response(amplitude profile plotted against the frequency) with and without thetransformer-coupled damping resistance.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 to FIG. 3 have been explained in the introduction in order toassist in understanding the problems which the present inventionaddresses, although it should again be mentioned that the recognition ofthe problem of system oscillations in a converter system as shown inFIG. 1, particularly with a supply network-side input inductor L_(K) inthe step-up converter mode in conjunction with a motor with a latticenetwork structure K, and the cause of that problem is not known from theprior art. Hence, the recognition of the problem represents an importantaspect of the present invention.

The system model of a converter system as shown in FIG. 1 will beanalyzed with regard to an effective path from the supply network to themotor star point. To this end, FIG. 4 shows the input-side supplyvoltage U_(N) with respect to ground, converted via the convertersystem, which has a first transfer function H₁(s), to the voltageU_(P600) on the positive intermediate circuit rail with respect toground. In the motor, voltage U_(P600) is converted via a secondtransfer function H₂(s) to a voltage U_(S) with respect to ground whichis present at the motor star point S.

In practice, a number of motors are frequently operated using oneconverter system, by a number of inverters W₁ to W₃ with motors M₁ to M₃connected to them being supplied from the intermediate circuit voltageU_(ZK), as shown in FIG. 5. The input converter E is supplied from thesupply network N via the filter arrangement F and supplies a number ofinverters W₁ to W₃, with motors M₁ to M₃ connected to them, from theintermediate circuit voltage U_(ZK).

Between the respective inverters W₁ to W₃ with connected motors M₁ toM₃, it must be remembered, with regard to system oscillations, thatthere is a natural system frequency f_(sys), which describes theresonant frequency f_(res)(sys) of the system, with N, F, E, W₁ to W₃ onthe converter system side. In contrast, the motors M₁ to M₃ have theirown resonant frequency f_(res), which corresponds to the naturalfrequency f_(res)(mot) of the respective motor.

The theoretical system analysis shown in FIG. 4 is thus carried outseparately for each respective motor, for which reason the transferfunction H₁(s) represents the effect of the filter F, the inductanceL_(K), the input converter E, all the inverters W, all the other motorsM and all the lines LT. Consequently, all of the elements in the dashedbox shown in FIG. 5 are modeled as H₁(s) when analyzing the behavior ofmotor M3.

A system oscillation which is excited in particular by the pulsing of asupply E and, to a lesser extent, also by the pulsing of the inverters Win the shaft modules, can be formed in such a converter or convertersystem. This pulsing results in periodic changing of the charge on theparasitic capacitances, as has been described with reference to FIG. 1.

If the supply network voltage U_(N) is regarded as an input variable,then this is mapped by the transfer function H₁(s) onto the outputvariable U_(P600) (when considering the positive intermediate circuitrail P600). Except for 600 V DC components, the voltage U_(P600) isapplied in the common mode to the motor terminals, and this correspondsto an unbalanced system or zero phase-sequence system.

The motor line LT can in theory be associated with both H₁(s) and H₂(s).In this case, as mentioned, the motor line LT will be associated withH₁(s). In the frequency band under consideration, the line LT can beregarded as an electrical short.

As already mentioned, the passive circuit formed in this way andillustrated in FIG. 2 has a natural resonant frequency f_(res)(sys) orf_(sys), at which this system starts to oscillate. Accordingly, inaddition to the voltage variations with an amplitude of 600 V, forexample, that occur during operation, an additional, undesirableoscillation with an amplitude of up to several hundred volts is alsomodulated onto the voltages on the intermediate circuit rails P600 andM600. This means that the output voltages from the inverter W withrespect to ground are no longer in step-function form, as is the casebetween two phases U, V, W, with the output voltages now including thesystem oscillations present on the intermediate circuit rails P600 andM600.

This is illustrated in FIG. 6, which shows a balanced drive for themotor M from the intermediate voltage circuit C_(ZK) on the basis of twophases L1 and L2, by way of example. FIG. 6 shows the intermediatecircuit with the intermediate circuit capacitance C_(ZK) and theintermediate circuit rails P600 and M600, from which, using a simplifiedinverter in the form of a bridge circuit with the switches S1 to S4, avoltage U_(L1L2) or a current i is produced in order to supply twosections L1 and L2 of the motor M, which are connected at the motor starpoint S, each having inductances L_(H). The motor has the previouslydescribed discharge capacitance C_(M) with respect to ground potential.

FIG. 7 shows the profile of the voltage U_(L1L2) between the phases L1and L2 plotted against time t and compared with the respective switchingstates of the switches S1 to S4 in the bridge of the inverter W, whichare plotted underneath, likewise with respect to time t. The switches S1and S2 represent the first bridge arm, and the switches S3 and S4represent the second bridge arm. In this case, switches in one phase arealways in opposite directions to one another since, otherwise, theintermediate circuit would be short-circuited.

The four states 1, 2, 3 and 4 are assumed in order to illustrate theswitching states of the two bridge arms, S1/S2 and S3/S4. In state 1,S1=0, S2=1 and S3=0, S4=1 with the state “−−” for the phases L1 and L2.Thus, in this case, what are referred to as zero vectors NZ are switchedand there is no voltage U_(L1L2) between the phases L1 and L2. In state2, S1=1, S2=0 and S3=0, S4=1. This results in the state “+−”, with avoltage U_(L1L2) of 600 V between the phases L1 and L2. In state 3,S1=1, S2=0 and S3=1, S4=0. This results in the state “++”, that is tosay zero vectors NZ are once again switched, and there is no voltageU_(L1L2) between the phases L1 and L2. Finally, in state 4, S1=0, S2=1and S3=1, S4=0. This results in the state “−+” with a voltage U_(L1L2)of −600 V between the phases L1 and L2. A new state 1 then starts, etc.

FIG. 8 also shows the voltage of phase L1 with respect to ground,plotted against time t for these states 1 to 4, that is to sayconsidered asymmetrically. In this case, the phenomenon described abovecan be seen, as a result of which the voltage profile does not have anideal step-function form since it is modulated with the undesirablesystem oscillations of the converter system from FIG. 1 and FIG. 4 withan amplitude, for example, of approximately 150 V. The same applies, insome circumstances, to a constant amplitude shift for the unbalancedvoltage of phase L2 with respect to ground, which is shown in FIG. 9. Ascan be seen, both phases L1 and L2, and thus also the intermediatecircuit rails P600 and M600, oscillate in time with one another. Thismeans that they are always shifted “in parallel”, that is to say withoutphase shift.

It is clear from this that the problem of resonant peaks is caused byunbalanced currents i. For this reason, it is worthwhile analyzing thearrangement as a common-mode system, a section of which is shown in FIG.10 in the form of an unbalanced drive for the motor M from theintermediate voltage circuit C_(ZK). In this case it is assumed that allthe motor phases U, V, W or L1 to L3 form an inductance L_(σ) which isgoverned by the motor winding and is terminated by the dischargecapacitance C_(M) to ground.

If one now considers the two phases L1 and L2 jointly, as a common-modesystem (referred to as L1&&L2 in the following text) then this resultsin the voltage with respect to ground over time shown in FIG. 11. Nocommon-mode signal can be shown in the states 2 and 4 in the common-modesystem for L1&&L2 from the “parallel” shift of the individual phases L1,L2 which can be seen in FIG. 8 and FIG. 9, since in this case the phasesL1 and L2 are at a different potential (in the sketched situation, theDC voltage difference is 600 volts). Since only two phases areconsidered, they produce an average of zero volts in the common mode. Inthe other states 1 and 3, the voltage of L1&&L2 over time corresponds tothat of L1 in FIG. 8, and L2 in FIG. 9.

The voltage of L1&&L2 over time as shown in FIG. 11 in the common-modesystem can in this case be separated into a fundamental GW and aharmonic OW. These are shown separately in FIG. 12 and FIG. 13. Theillustration in FIG. 12 shows the voltage of the fundamental GW overtime. This clearly shows that this voltage profile describes the desiredstep-function form switching states with −300 V in state 1, 0 V instates 2 and 4 owing to the “parallel” shift, and +300 V in state 3. Theharmonic OW of the voltage L1&&L2, shown over time in FIG. 13, describesan essentially constant sinusoidal profile with an amplitude of, forexample, 150 V.

The harmonic or system oscillation is thus applied to the motor M in allstates 1 to 4, as a result of which this phase-ground tuned circuit isalways excited in the motor M, as shown in FIG. 2. If this systemoscillation is now in the vicinity of a natural motor frequency, and/orthe motor M has a pronounced resonance in the vicinity of the frequencyof the system oscillation, undesirable resonant peaks can occur. A“maximum” oscillation amplitude of this phase-ground tuned circuit isgenerally prevented only by the discontinuities in the harmonicresulting from the switching from one state to the next.

With regard to the theoretical system analysis of the problems shown inFIG. 4, as mentioned above, the amplitude of such a system oscillationf_(sys) in this case depends essentially on two factors. Namely, theintrinsic damping in the system, which is inversely proportional to theQ-factor of the tuned circuit with the damping increasing as thefrequency rises, and the excitation, that is to say the nature of thesupply (for example diode supply or regulated supply) and the magnitudeof the intermediate circuit voltage U_(ZK). Particularly pronouncednatural system oscillations can thus be observed in converter systemswhich have a large number of shaft modules W and motors M, and longmotor lines LT. The frequency band of the natural system oscillationsf_(sys) in this case generally extends from approximately 10 kHz forlarge converter systems up to more than 50 kHz for smaller convertersystems.

The amplitude and frequency thus depend on the configuration and theextent of the system, for example:

the nature of the supply E (regulated or unregulated);

the number of shafts or motors M which are operated using one convertersystem UR; and

the length of the power lines LT.

It should thus be stated at this point that converter systems with anintermediate voltage circuit may have natural oscillations on theintermediate circuit rails P600, M600 with respect to ground. These areparticularly pronounced in multi-shaft systems and with a regulatedinput converter E, particularly when used in the step-up converter mode.The motor M in an unbalanced system is thus excited virtually at asingle frequency irrespective of the pulse patterns on the individualphases U, V, W or L1 to L3. The transfer function H₂(s) results in thisexcitation being mapped onto the output side, namely the voltage U_(S)at the star point S with respect to ground.

All electric motors M, irrespective of the type, qualitatively have atransfer function H₂(s) with respect to ground whose amplitude/frequencyresponse A(f) is as shown in the illustration in FIG. 14. This has apronounced resonant frequency f_(res)(mot) or f_(res). The transferfunction H₂(s) can in this case be described as:

H ₂(s)=U _(P600) /U _(S)

The frequency of the pronounced resonant peak of the motor depends onthe inductive and capacitive elements in the motor with respect toground, and is thus governed by:$f_{res} \propto \frac{1}{\sqrt{L_{M} \cdot C_{M}}}$

where L_(M)=f(L_(PE)) is the effective inductance and C_(M)=f(C_(PE)) isthe effective capacitance of the motor M with respect to groundpotential PE or in the zero phase-sequence system. The precise functionsin this case depend on the respective measurement method and on theequivalent circuits used.

If there are a number of star points S, identical tuned circuits areconnected in parallel. The capacitance per tuned circuit is thengoverned by: $\overset{\sim}{C} \propto {\frac{C_{M}}{{number}_{S}}.}$

The inductance depends on the number of series-connected coils, withthere being a number of star points S, particularly when using fieldcoil technology. Since the individual coils can be regarded as beingmagnetically decoupled from one another, it can furthermore be statedthat: $\overset{\sim}{L} \propto \frac{n_{1S}}{{number}_{S}}$

where n_(1s) is the number of coil assemblies for one star point S, andnumber_(s) is the number of star points S.

Thus, for motors of the same physical size but in which identical coilgroups are connected differently:$f_{res} \propto \frac{1}{\sqrt{\frac{1}{{number}_{S}} \cdot \frac{1}{{number}_{S}}}} \propto {{number}_{S}.}$

The influence of the motor size on the resonant frequency f_(res) can beestimated as follows: $C = \frac{ɛ \cdot A}{d}$

where

Aslot areaD·LG

where D is the diameter and LG the length of the motor.

The influence of the motor size with the characteristics otherwise beingconstant is thus reflected by:$f_{res} \propto \frac{1}{\sqrt{{slot}\quad {area}}} \propto {\frac{1}{\sqrt{D \cdot {LG}}}.}$

Ignoring the natural damping, which increases as the frequency f rises,resulting from eddy current losses, hysteresis etc. and in particular ifthe motor M is regarded as a lattice network K, as appears to bemacroscopically plausible particularly in the case of motors using fieldcoil technology since the coil groups are connected in series, thisresults in the amplitude/frequency response A(f) shown in FIG. 15. Theresponse graph A(f) has a number of local maxima which describe a numberof resonant frequencies f_(res 1) to f_(res n), with the first resonantpeak f_(res 1), which occurs at the lowest frequency, being dominant andthus representing the governing or pronounced resonant frequencyf_(res).

There are thus frequencies, particularly the lowest resonant frequencyf_(res), at which considerably greater voltages occur at the motor starpoint S than at the input terminals of the motor M and these are, forexample, greater by a factor of 3 to 4. In this case, it can be statedthat the resonant peak becomes higher the lower f_(res) is.Geometrically large torque motors are thus particularly at risk, inwhich resonant points f_(res) which are in the vicinity of or areprecisely at the frequency f_(sys) of the natural system oscillationscan be formed particularly easily due to the slot area and the number ofstar points S.

This knowledge is important in considering the lattice network structureK as shown in FIG. 3. This is because such a lattice network structure Kcan be regarded not only macroscopically in motors using field coiltechnology, but also, in principle, with other types. To this end, themotor M together with its motor winding can also be regarded in entirelygeneral form as a microscopic lattice network K composed of identicalfour-pole networks V1 to Vn, as is shown in the illustration in FIG. 16.Each four-pole network V1 to Vn in this case comprises an inductance L,which is connected in series with a non-reactive resistance R. Theoutput voltage is in this case dropped in parallel with a capacitance C,which is connected as a voltage divider with L and R.

In order to illustrate this structure, the lattice network structure Kwill be described in more detail. To this end, FIGS. 17 to 19 show acomparison of the construction of a motor winding using field coiltechnology on the basis of its unbalanced electrical equivalent circuit.The illustration in FIG. 17 shows a coil group in the phase U. Each coilgroup of the motor winding MW, comprising a number of series-connectedfield coils PS1 to PS3, forms with respect to ground a macroscopic LClattice network with an inductance L and a capacitance C. The start andend of this lattice network K are the input terminal U and star point Sof the motor M. As described above, this lattice network K has a numberof resonant frequencies f_(res).

If such a lattice network is excited at the input U (phase to ground)for example sinusoidally at its lowest natural frequency f_(res), whichis generally the most pronounced, then it can supply a considerablyhigher voltage at the star point S than at the input U. In the worstcase, this voltage can lead to a breakdown in the main insulation in thevicinity of the star point S. Such sinusoidal excitation can occur inparticular as a result of inadvertent system oscillation f_(sys) (asdescribed above) of the entire converter system.

The described mechanism is most clearly pronounced when at rest since,in this case, all the phases U, V, W or L1, L2, L3 are switched at thesame time. The natural frequencies and damping levels in the latticenetwork K depend on the construction of the motor winding, for example,on:

the number of coils per path

the number of turns (inductance)

the shape of the coils (capacitance)

the encapsulation (capacitance).

The illustration in FIG. 18 shows how the unbalanced equivalent circuitof such an arrangement as shown in FIG. 19 is obtained from an exampleof a winding design as shown in FIG. 17 and the installed position ofthe field coils PS1 to PS3 in the laminated core B.

The winding organization means that the first layer (solid circles) of afield coil PS1 to PS3 always has a larger capacitance C with respect tothe laminated core B (ground potential) than the other layers (hollowcircles).

The respective inductance L is formed by the respective field coil PSI,PS2, PS3 itself, on the assumption that the mutual inductance betweenthe coils can be ignored since, at the frequencies under consideration(for example 20 kHz), the iron only slightly amplifies and guides themagnetic flux. This is confirmed by the fact that the omission of thesecondary part scarcely changes the measured values (frequency,amplitude).

The illustrated structure thus has as large a number of lattice networkelements (n=3) as field coils PS1 to PS3. If a winding section comprisesm parallel-connected paths (lattice structures), then m=3 paths areconnected in parallel for operation with a zero vector NZ (simultaneousswitching of all the input terminals) for the phases U, V, W.

The natural damping in the system is not shown in FIG. 19. Initially,this should be regarded as the non-reactive resistance R of eachinductance L.

In the following example, the transfer function of the lattice networkstructure which is shown in FIG. 16 and comprises the four-pole networksV1 to Vn from the phase terminals to the star point S is calculated fora motor M by using the variables L_(Z), C_(M), m and n.

In this case, C_(M) describes the winding-ground capacitance, and L_(Z),R_(Z) describe the impedance of one winding path. The expression windingpath in this case means the series-connected coils from one phaseterminal U, V or W to a star point S, with any parallel connectionswhich exist within a winding section being disconnected.

This allows the parameters for a lattice network element or four-polenetwork to be determined as follows: $L = \frac{L_{Z}}{n}$$C = \frac{C_{M}}{3 \cdot n \cdot m}$

m: number of parallel paths;

n: number of coils in series.

The loss resistance R of the inductance L is initially set to$R = \frac{R_{Z}}{n}$

with the value for R_(Z) in the series model being predetermined, forexample, 10 kHz.

This value can be only a first approximation, since R is highlyfrequency-dependent and is determined at a lower frequency than thefrequencies which actually occur.

The Z-matrix of an individual four-pole element Z_(v) is:${\underset{\_}{Z}}_{v} = \begin{bmatrix}{R + {sL} + \frac{1}{sC}} & \frac{1}{sC} \\\frac{1}{sC} & \frac{1}{sC}\end{bmatrix}$

The lattice matrix A_(v) can be formed as follows:${\underset{\_}{A}}_{v} = \begin{bmatrix}{{s^{2}{LC}} + {sRC} + 1} & {{sL} + R} \\{sC} & 1\end{bmatrix}$

The A-matrix for the entire lattice network K is then:

A _(tot)=A_(v) ^(n)

with the element${{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack} = \frac{{\underset{\_}{U}}_{1}}{{\underset{\_}{U}}_{2}}$

The amplitude/frequency response A(s) is thus, as a complexgeneralization of A(f):${A(s)} = {20\quad 1\quad {g\left( {\frac{1}{{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack}} \right)}\quad {in}\quad {db}}$

If it is assumed that the damping resistance R increases with frequency,then the higher resonant frequencies would be even more strongly damped.

If the A-matrix for the entire lattice network K is used to form theZ-matrix, then the element Z_(in) is equal to the input impedance of thelattice network.

This results in the input impedance, as follows:$Z_{in} = {{\frac{{\underset{\_}{Z}}_{tot}\left\lbrack {1,1} \right\rbrack}{m \cdot n}} = {\frac{1}{m \cdot n} \cdot {\frac{{\underset{\_}{A}}_{tot}\left\lbrack {1,1} \right\rbrack}{{\underset{\_}{A}}_{tot}\left\lbrack {2,1} \right\rbrack}}}}$

The illustration in FIG. 20 shows a lattice network structure K formedin a motor M, as has been explained with reference to FIG. 3, but nowwith an impedance Z according to the invention which istransformer-coupled into all three phases U, V, W of the motor M. Thisis done, for example, by routing each phase U, V, W of the motor M,whose winding section represents a lattice network K, through one andthe same magnetic core MK, which is either sufficiently lossy owing toits material characteristics or else has a winding which is electricallyshort-circuited by an impedance Z (as in FIG. 20). Since all the phasesare routed through one magnetic core MK, only the unwanted common-modeprocesses are damped with respect to the natural system oscillations inthe converter system.

FIG. 21 shows the transformer equivalent circuit relating to thearrangement shown in FIG. 20. The magnetic core MK is in this casedescribed by the four-pole network equivalent circuit of a transformerwith the main inductance of the transformer and the total inductance ofthe coupling coil L_(H1), together with the stray inductances L_(σ1) andL_(σ2). The impedance Z is in this case composed of a series circuitformed by a non-reactive damping resistance R_(a) and a parasiticinductance L_(R), which is in general formed by the supply lines of theresistance. The impedance Z may, of course, also include other elementssuch as capacitances, which are used to compensate for L_(σ2) and L_(R).

On the basis of this equivalent circuit for the coupling core MK and theimpedance Z, R_(a) and L_(H1) can now be dimensioned so that asufficiently large time constant is achieved resulting in a satisfactorydamping effect with minimized losses in the impedance Z. To ensure thisresult, the time constant should correspond to approximately half thelowest motor resonant frequency f_(res).

The resistive component R_(a) is preferably dimensioned in accordancewith$R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}$

In this case, L describes the inductance and C the discharge capacitanceof a four-pole network V1 . . . Vn, or of a lattice network LC elementas illustrated in FIG. 16 or FIG. 19.

The main inductance of the transformer and the total inductance of thecoupling core MK are preferably dimensioned in accordance with${L_{Hl}>=\frac{R}{2\pi \quad f_{0}}},{where}$${f_{0}<={\frac{1}{2} \cdot f_{res}}},$

and f_(res) is the pronounced resonant frequency of the motor M.

It can be seen that the inductance L_(H1) should not be reduced bysaturation effects. For this reason, the coupling core MK should bedimensioned such that it cannot enter saturation at any operatingpoints.

FIG. 22 shows the amplitude/frequency response A(f) with respect toground potential with and without the impedance Z beingtransformer-coupled according to the invention into all three phases U,V, W of the motor M. The solid line corresponds to the undampedsituation as shown in FIG. 15, with the first resonant frequency as themost pronounced resonant frequency f_(res). Transformer damping of theunwanted common-mode processes, dimensioned as described above, resultsin the dotted profile, where the resonant peak at f_(res) isconsiderably lower. Thus, if this is located in the vicinity of thenatural system frequency f_(sys) of the converter system, then there isno need to be concerned about resonant peaks with the described negativeconsequences.

The advantages of the invention are a reduction in the load on theinsulation system with respect to ground, which improves the reliabilityand robustness of the motor and elimination of the need for expensiveadditional motor insulation that reduces the rating of the motor, sincethe voltage load remains in areas which are regarded as capable ofhandling the voltage load using standard materials based on the presentprior art for low-voltage motors.

The foregoing merely illustrates the principles of the invention inexemplary embodiments. Various modifications and alterations to thedescribed embodiments will be apparent to those skilled in the art inview of the teachings herein. For example, the principles of theinvention could be applied if there are a number of motor star points S,as is the situation, for example, in linear motors or torque motors. Tothis end, a suitable impedance Z for each motor star point S is in eachcase transformer-coupled into all the motor phases U, V, W leading to anindividual motor star point S. In principle, resonant peaks also existin motors using what is referred to as wild winding (standard forlow-voltage motors), so that the principles of the invention could beused for these motors and for motors other than the field coiltechnology motors chosen for illustrative purposes. It will thus befully appreciated that those skilled in the art will be able to devisenumerous systems and methods which, although not explicitly shown ordescribed, embody the principles of the invention and thus are withinthe spirit and scope of the invention as defined in the appended claims.

We claim:
 1. A method for damping resonant electrical peaks at a motorstar point in an electric motor which is operated using an intermediatevoltage circuit converter with an input-side inductance, said input-sideinductance, owing to characteristics of the winding sections of saidelectric motor, has a frequency response with at least one resonantfrequency with respect to ground potential, comprising: introducing intoall phase paths leading to at least one motor star point, an impedancefor damping capacitive discharge currents to ground potential, which arecaused by system oscillations excited asymmetrically with respect toground in the motor phases of the converter in the winding sections,further wherein all the motor phase paths leading to at least one motorstar point-are routed through a lossy magnetic coupling; wherein thecoupling core has a winding that is short-circuited via an impedance. 2.The method according to claim 1, wherein said lossy coupling core iscoupled to the input to the motor.
 3. The method according to claim 1,wherein the impedance comprises a non-reactive resistance.
 4. The methodaccording to claim 3, wherein each winding section of the motor forms anLC lattice network, and the non-reactive resistance, R_(a) isdimensioned in accordance with the formula:${R_{a} \geq {\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}},$

where L is the inductance and C the discharge capacitance of one latticenetwork element in the LC lattice network structure.
 5. The methodaccording to claim 3, wherein the total inductance, L_(H1) of a couplingcircuit formed with the coupling core is dimensioned in accordance withthe formula: $\begin{matrix}{{{L_{H1}\quad>=\quad \frac{R}{2\quad \pi \quad f_{0}}},}\quad} \\\text{wherein R is non-reactive resistance, and where, preferably,}\end{matrix}$ ${f_{0}\quad<=\quad {\frac{1}{2} \cdot f_{res}}},$

with f_(res) being a pronounced resonant frequency.
 6. The methodaccording to claim 1, wherein the converter is operated together withthe input-side inductance in order to provide a supply based on thestep-up converter principle.
 7. The method according to claim 1, whereinthe input-side inductance comprises an inductor.
 8. An electric motorfor operation using an intermediate voltage circuit converter having aninput-side inductance and having a frequency response that is governedby winding inductances and discharge capacitances, with a pronouncedresonance with respect to ground potential, in which all the motor phaselines leading to a motor star point are routed through a lossy magneticcoupling core, wherein the coupling core has a winding which isshort-circuited via an impedance.
 9. The electric motor according toclaim 8, wherein the coupling core is coupled to at the input to themotor.
 10. The electric motor according to claim 8, wherein theimpedance is so pronounced that it damps capacitive discharge currentsto ground potential, which are caused by system oscillations excitedasymmetrically with respect to ground in the motor phases of a convertersystem in winding sections of the motor.
 11. The electric motoraccording to claim 8, wherein the impedance comprises a non-reactiveresistance.
 12. The electric motor according to claim 11, wherein eachwinding section of the motor forms an LC lattice network, and thenon-reactive resistance, R_(a), is dimensioned in accordance with theformula:${R_{a}>={\frac{1}{2} \cdot \frac{1}{3} \cdot \sqrt{\frac{L}{C}}}},$

where L is the inductance and C the discharge capacitance of one latticenetwork element in the LC lattice network structure.
 13. The electricmotor according to claim 12, wherein the impedance is so pronounced thatit damps common-mode currents excited asymmetrically with respect toground in the motor phases of a converter system in the lattice networkstructure.
 14. The electric motor according to claim 11, wherein thetotal inductance L_(H1), of a coupling circuit formed with the couplingcore is dimensioned in accordance with the formula:${L_{H1} \geq \frac{R}{2\pi \quad f_{0}}},{{where}\quad {preferably}},{f_{0} \leq {\frac{1}{2} \cdot f_{res}}},$

with f_(res) being a pronounced resonant frequency.
 15. The electricmotor according to claim 8, wherein the coupling core is dimensioned insuch a manner that it does not enter saturation at any operating pointof the motor.
 16. The electric motor according to claim 8, furthercomprising winding sections using field coil technology, wherein eachwinding section form a lattice network structure composed of inductancesand discharge capacitances, in which case the impedance is used fortransformer damping of these lattice network structures.
 17. An electricmotor according to claim 8, wherein said motor employs wild windingtechnology which has low resonant frequencies by virtue of itsconstruction.